Switched-capacitor power converters

ABSTRACT

An apparatus for providing electric power to a load includes a power converter that accepts electric power in a first form and provides electric power in a second form. The power converter comprises a control system, a first stage, and a second stage in series. The first stage accepts electric power in the first form. The control system controls operation of the first and second stage. The first stage is either a switching network or a regulating network. The second stage is a regulating circuit when the first stage is a switching network, and a switching network otherwise.

CROSS REFERENCE TO RELATED APPLICATION

This application claims the benefit of the priority date of U.S.Provisional Application No. 62/189,909, filed on Jul. 8, 2015, thecontents of which are hereby incorporated by reference in theirentirety.

FIELD OF INVENTION

This invention relates to power converters, and in particular, to dc-dcpower converters.

BACKGROUND

It is known in the art that electrical devices require electric power tooperate. However, some electrical devices are more omnivorous thanothers. For example, a tungsten filament light bulb will operate over awide range of voltages. Although it may be dim at low voltages, andalthough it may burn out prematurely at high voltages, it does notsimply stop operating.

Digital circuits, however, are more finicky in their requirements. Adigital circuit demands power with particular characteristics. Aprocessor that receives power falling short of these characteristicswill not just compute more slowly. It will simply shut down.

Unfortunately, power is not always delivered in a form that amicroprocessor-based system will find acceptable. For example, in ahandheld device, the battery voltage ranges from fully charged to almostzero. Thus, most such systems require something that accepts power inraw form and delivers it to the system in a form that the system findsmore palatable.

This critical but unglamorous task falls upon the power converter.

A variety of power converters are known. These include power convertersdescribed in U.S. Pat. Nos. 8,860,396, 8,743,553, 8,723,491, 8,503,203,8,693,224, 8,724,353, 8,339,184, 8,619,445, 8,817,501, U.S. Patent Publ.2015/0077175, and U.S. Pat. No. 9,041,459.

The contents of all the foregoing patents are herein incorporated byreference.

SUMMARY

In one aspect, the invention features an apparatus for providingelectric power to a load. Such an apparatus includes a power converterthat accepts electric power in a first form and provides electric powerin a second form. The power converter includes a control system, andfirst and second stages in series. The first stage accepts electricpower in the first form. The control system controls operation of thefirst and second stages. The first stage is either a switching networkor a regulating network. The second stage is a regulating network whenthe first stage is a switching network. On the other hand, the secondstage is a switching network when the first stage is a regulatingnetwork.

Among the embodiments are those in which the control system controls atleast in part based on a voltage measured between the first and secondstages.

Also among the embodiments are those in which the first stage is aregulating network, those in which the first stage is a switchingnetwork, and those in which it is the second stage that is a switchingnetwork, such as a cascade multiplier. In either case, the switchingnetwork can be a cascade multiplier.

In some embodiments, at least one of the stages includes a switchingnetwork having first and second terminals. Among these are embodimentsin which these terminals are isolated, embodiments in which they have acommon ground, and embodiments in which they have separate grounds.

In other embodiments, at least one of the stages includes a switchingnetwork having a first and second switching circuits, each of which hasfirst and second terminals. In these embodiments, the first terminal ofthe second switching circuit connects to the second terminal of thefirst switching circuit. Among these are embodiments in which the twoswitching circuits have different voltage-transformation ratios, andembodiments in which they have the same voltage-transformation ratios.

In some embodiments, the switching network includes first and secondswitching circuits in series, whereas in others, it includes first andsecond switching circuits in series-parallel.

Some embodiments of the power converter further include a third stage inseries with the first stage and the second stage so that the secondstage is between the first stage and the third stage. These embodimentsinclude those in which the first and third stages are switchingnetworks, those in which the first and third stages are regulatingnetworks, and those in which the third stage is operated with a dutycycle that causes the third stage to become a magnetic filter.

In some embodiments, the switching network includes a cascademultiplier. Among these are those embodiments in which the cascademultiplier is a single-phase cascade multiplier, those in which it isasymmetric, those in which it is a step-down, multiplier, and any thosein which it is any combination thereof. Also among these embodiments arethose in which the cascade multiplier is a dual-phase cascademultiplier. In this case, the cascade multiplier could be a symmetriccascade multiplier, or one that includes parallel pumped capacitors, orone that lacks DC capacitors.

In some of the foregoing embodiments, the cascade multiplier creates anauxiliary voltage to drive an additional circuit. Among these areembodiments that include a level shifter connected to be driven by theauxiliary voltage, and those in which a gate driver is connected to bedriven by the auxiliary voltage.

In some embodiments, the switching network includes first and seconddual-phase cascade multipliers, and a phase node shared by both cascademultipliers. In these embodiments, the first cascade multiplier, whichis stacked on the second, is asynchronous and the second cascademultiplier is synchronous. Among these embodiments are those in whichthe first and second cascade multipliers operate at the same frequency,and those in which the first and second cascade multipliers operate atdifferent frequencies.

In some embodiments, the regulating network includes a buck converter.Among these are embodiments in which the buck converter includes firstand second terminals with the same reference voltage. Examples includethose in which the buck converter's first and second terminals are atdifferent reference voltages, those in which the buck converter hasthree terminals, and those in which the buck converter has a floatingnode at a floating voltage. In embodiments that have a floating node,the floating node can be between two loads or between two sources.

A variety of other regulating networks are contemplated. These include abuck-boost converter, a boost converter, and even a four-terminalnon-inverting buck-boost converter.

In some embodiments that use a boost converter as a regulating network,the switching network includes a step-down single-phase asymmetriccascade multiplier. In some of these embodiments, selection switchesconnected to the regulating network cause the switching network tooutput a fraction of its normal output voltage. In others, switches areoriented so that cathodes of parasitic diodes corresponding to theswitches connect to each other. Among these embodiments are those inwhich the first stage is a regulating network.

Embodiments include those in which those in which the regulating networkregulates plural wires, those in which it regulates at most one wire,and those in which it regulates a particular one of plural wires basedon an input voltage to the regulating network.

Also among the embodiments are those in which the regulating network hasplural output ports and those in which it is a multi-tap boostconverter. Among these are embodiments in which the switching networkincludes a single-phase step-down switched-capacitor circuit.

In yet other embodiments, the power converter floats above ground.

In some embodiments, the switching network is reconfigurable. In otherembodiments, it is the regulating network that is reconfigurable. In yetothers, both are reconfigurable. In either case, there are embodimentsin which a magnetic filter connects to whichever of the two arereconfigurable. Thus, a magnetic filter could be connected to either thereconfigurable switching network or the reconfigurable regulatingnetwork.

In some embodiments, the switching network includes a dual-phaseswitched capacitor circuit. Among these are embodiments in which theswitched capacitor circuit includes pump capacitors in series and DCcapacitors in series.

In some embodiments, the switching network includes a dual-phaseswitching circuit including DC capacitors that store charge from theregulating network only during a dead-time transition during which theswitching network is between states.

In yet other embodiments, the regulating network includes an inductorthat promotes adiabatic charge transfer within the switching network.

Some embodiments also include a magnetic filter connected to theswitching network to promote adiabatic charge transfer within theswitching network. Among these are embodiments in which the magneticfilter is connected between the switching network and a load, those inwhich the magnetic filter is connected between the switching network anda source, and those in which the regulating network and the magneticfilter cooperate to promote adiabatic charge transfer within theswitching network.

Embodiments further include those that have a circuit connected to theswitching network to constrain current flow out of the switchingnetwork, and those that have a circuit connected to the switchingnetwork to promote adiabatic charge transfer within the switchingnetwork.

In some embodiments, the switching network includes a two-phasestep-down switching network and the regulating network is a step-downnetwork. Among these are embodiments in which the switching networkincludes a cascade multiplier. In those embodiments that include acascade multiplier, the regulating network can include a buck converter.Also among these embodiments are those in which regulating networkpromotes adiabatic charge transfer.

In still other embodiments, the switching network includes a step-downsingle-phase asymmetric cascade multiplier and the regulating networkincludes a converter that causes voltage to step down. In some of theseembodiments, it is the first stage that is a switching network.

In some embodiments, the regulating network includes a multiple-tap buckconverter configured to have two operating modes. Among these areembodiments in which the switching network provides first and secondvoltage rails that, in operation, are maintained at different voltages.

Yet other embodiments are those in which the regulating network includesa buck converter having multiple taps and configured have threeoperating modes. Among these are embodiments in which the switchingnetwork provides first, second, and third voltage rails that, inoperation, are maintained at different voltages.

Other embodiments of the apparatus are those in which the switchingnetwork includes a two-phase switched-capacitor circuit and theregulating network is a buck converter.

Also among the embodiments are those in which the regulating networkincludes parallel first and second regulating circuits.

In some embodiments, the power converter includes first and secondoutputs. In operation, the first output and second outputs beingmaintained at corresponding first and second voltage differences. Thefirst voltage different is a difference between a first voltage and asecond voltage, and the second voltage difference is a differencebetween a third voltage and the second voltage.

In some embodiments, the regulating network includes first, second, andthird regulating circuits in parallel.

In other embodiments, the power converter includes first second, andthird outputs. In operation, the first, second, and third outputs aremaintained at corresponding first second, and third voltage differences.The first voltage different is a difference between a first voltage anda second voltage. The second voltage difference is a difference betweena third voltage and the second voltage. And the third voltage differenceis a difference between a fourth voltage and the second voltage.

In some embodiments, the power converter has a first terminal and asecond terminal such that, in operation, a first voltage different ismaintained across the first terminal and a second voltage difference ismaintained across the second terminal. The first voltage difference is adifference between a first voltage and a second voltage, and the secondvoltage difference is a difference between a third voltage and thesecond voltage, with the second voltage being variable. Some of theseembodiments also have a third stage that provides the second voltage.Also among these are embodiments in which the third stage includes aswitched-mode power converter, a switched capacitor converter, a buckconverter, or a cascade multiplier.

In some embodiments, the power converter is configured to provide ACoutput with a non-zero DC offset.

In other embodiments, the switching network includes a reconfigurableasynchronous cascade multiplier, and the regulating network is connectedto the switching network to enable the switching network to cause eithera step up in voltage or a step down in voltage. In some cases, the theregulating network includes a four-switch buck-boost converter.

In still other embodiments, the first stage is a switching network thatincludes a reconfigurable cascade multiplier that operates synchronouslyin a single-phase, and the regulating network includes a four switchbuck boost converter. Among these are embodiments in which theregulating network connects to the switching network at a point thatenables the switching network to step voltage up or step voltage down.

Embodiments also include those in which the switching network includes acascade multiplier with a charge pump embedded therein. The charge pumpcan have a variety of characteristics. For example, the charge pump canbe reconfigurable, or it can be a fractional charge pump. Alternatively,the embedded charge pump operates in multiple modes, each of whichcorresponds to a voltage transformation ratio. Or the cascade multipliermight include a reconfigurable two-phase asynchronous step-down cascademultiplier. In any of these embodiments, the regulating network couldinclude a two-phase boost converter.

In still other embodiments, the power converter further includes a thirdstage in series with the first stage and the second stage, wherein thesecond stage is between the first stage and the third stage, both ofwhich are switching networks. The regulating circuit includes a buckconverter, and both switching networks include a single-phaseasynchronous step-up cascade multiplier. These embodiments include thosein that further include a stabilizing capacitor at an output of theregulating network.

In still other embodiments, the power converter further includes a thirdstage in series with the first stage and the second stage, with thesecond stage being between the first stage and the third stage. In theseembodiments, the first stage and the third stage are switching networks,the regulating circuit includes a buck-boost converter, the firstswitching network includes a single-phase asynchronous step-up cascademultiplier, and the second switching network includes a single-phasesynchronous step-up cascade multiplier. Among these embodiments arethose that also have a stabilizing capacitor at an output of theregulating network.

In some embodiments, the power converter further includes a third stagein series with the first and second stage, with the second stage beingbetween the first stage and the third stage. The first and third stageare both regulating networks. However, the first stage includes a boostconverter, and the third stage includes a buck converter. The switchingnetwork includes first and second cascade multipliers having equalnumbers of stages. Some of these embodiments also have a phase pumpshared by the first and second cascade multipliers. In others, the firstand second cascade multipliers operate 180 degrees out of phase. And inyet others, the cascade multipliers comprise corresponding first andsecond switch stacks, and an output of the switching network is avoltage difference between a top of the first switch stack and a top ofthe second switch stack.

In some embodiments, the power converter further includes a third stagein series with the first and second stages, with the second stage beingbetween the first stage and the third stage. In these embodiments, thefirst stage and the third stage are regulating networks, the first stageincludes a three-level boost converter, the third stage includes a buckconverter, and the switching network includes first and second cascademultipliers having unequal numbers of stages.

In other embodiments, the switching network receives current that has afirst portion and a second portion, wherein the first portion comes fromthe regulating network, and the second portion, which is greater thanthe first, bypasses the regulating network.

In some embodiments, the power converter further includes a third stagein series with the first stage and the second stage, the second stagebeing between the first stage and the third stage. The first stage is afirst regulating network, the third stage is a second regulatingnetwork, and the first stage includes a boost converter. The third stageincludes a buck converter. The switching network includes cascademultipliers having unequal numbers of stages. Among these embodimentsare those in which the second stage includes an additional inductorconnected to the first stage.

Yet other embodiments include a third stage. In these embodiments, thefirst stage includes a regulating network, the third stage includes aregulating network, the power converter provides a load with a firstvoltage difference, the first stage provides a second voltage differenceto the second stage, the second stage provides a third voltagedifference to the third stage, the first voltage difference is a voltagedifference between a first voltage and a second voltage, the secondvoltage difference is a voltage difference between a third voltage and afourth voltage, the third voltage difference is a voltage differencebetween a fifth voltage and a sixth voltage, the fourth voltage differsfrom the second voltage, and the sixth voltage differs from the secondvoltage. Among these embodiments are those in which the second stageincludes a reconfigurable switching network.

In some embodiments, the first stage includes a switching network havinga reconfigurable dual phase cascade multiplier with an embedded inductorconfigured to promote adiabatic charge transfer between capacitors inthe cascade multiplier. In some embodiments, the inductor is embedded ata location through which a constant current passes. Also among theseembodiments are those in which the second stage includes a zetaconverter, and those in which the cascade multiplier includes a phasepump with the inductor being embedded therein. In others of theseembodiments, the cascade multiplier includes pump capacitors, and theinductor is embedded at a location that maximizes the number of pathsthat pass between the inductor and a pump capacitor.

In some embodiments, the switching network includes a dual-phase cascademultiplier with pump capacitors in series. Among these are embodimentsin which the switching network has a variable transfer function andembodiments in which the switching network includes a phase pump thatincludes an embedded charge pump. In this latter case, the embeddedcharge pump includes switch sets, pump capacitors, and a controller thatoperates the switch sets to cause transitions between a first operatingmode and a second operating mode, each of which corresponds to atransfer function for the cascade multiplier. Among these cases arethose in which the controller operates the switch sets so that theembedded controller causes the cascade multiplier to have a transferfunction in which the cascade multiplier provides either a voltage gainor a voltage attenuation.

In another aspect, the invention features an apparatus for providingelectric power to a load includes a power converter that acceptselectric power in a first form and provides electric power in a secondform. The power converter includes a control system, a first stage, anda second stage in series. The first stage accepts electric power in thefirst form. The control system controls operation of the first andsecond stage. The first stage is either a switching network or aregulating network. The second stage is a regulating circuit when thefirst stage is a switching network, and a switching network otherwise.

These and other features will be apparent from the following detaileddescription and the accompanying figures, in which:

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a power converter having a regulating circuit and aswitching network in series;

FIG. 2 shows a power converter that is the converse of that shown inFIG. 1 ;

FIG. 3 shows a power converter having a switching network with twoswitching circuits in series;

FIG. 4 shows a power converter having a switching network with twoswitching circuits in series-parallel;

FIG. 5 shows a switching network between two regulating networks;

FIG. 6 shows a regulating network between two switching networks;

FIG. 7 shows an equivalent circuit for a switching network in which theinput and output are isolated;

FIG. 8 shows an equivalent circuit for a switching network in which theinput and output have a common ground;

FIG. 9 shown an equivalent circuit for a switching network in which theinput and output have a non-common ground;

FIG. 10 shows a first implementation of the switching network in FIG. 1;

FIG. 11 shows a second implementation of the switching network in FIG. 1in which the switching network is a dual-phase symmetric cascademultiplier having parallel pumped capacitors;

FIG. 12 shows a third implementation of the switching network in FIG. 1in which the switching network has a dual-phase asymmetric cascademultiplier stacked on top of dual-phase symmetric cascade multiplierwith shared phase nodes;

FIG. 13A shows a variant of the power converter in FIG. 10 ;

FIG. 13B shows a power converter similar to that shown in FIG. 13A, butwith the order of the regulating network and the switching networkhaving been reversed;

FIG. 14A shows a second implementation of the switching network in FIG.1 ;

FIGS. 14B and 14C show implementations of the phase pump shown in FIG.14A;

FIG. 15A shows a buck converter used to implement a regulating network;

FIGS. 15B-15C show non-isolating variants of the buck converter of FIG.15A;

FIGS. 16A-16B show non-isolating buck-boost converters used to implementa regulating network in any of the foregoing power converters;

FIGS. 17A-17B show non-isolating boost converters used to implement aregulating network in any of the foregoing power converters;

FIGS. 18A-18B show four-terminal non-inverting buck-boost converters;

FIG. 19 shows the power converter of FIG. 1 with a reconfiguringswitching network;

FIG. 20 shows the power converter of FIG. 1 with a reconfiguringregulating network;

FIG. 21 shows a power converter in which the regulating circuit has morethan two output ports;

FIG. 22 shows a power converter similar to that shown in FIG. 21 , butwith a dual-phase switched-capacitor circuit instead of as asingle-phase switched-capacitor circuit;

FIG. 23 shows a power converter with a magnetic filter between aswitching network and a load;

FIG. 24 shows a power converter with a magnetic filter between aswitching network and a power source;

FIG. 25 shows the power converter of FIG. 22 modified to include amagnetic filter between the switching network and the load;

FIG. 26 shows a power converter that uses a two-phase step-downswitching network connected in series with a step-down regulatingnetwork;

FIG. 27 shows a power converter similar to that shown in FIG. 19 butwith the order of the switching network and the regulating networkreversed;

FIG. 28 shows a power converter in which the regulating network is abuck converter with multiple taps;

FIG. 29 shows a power converter similar to that shown in FIG. 28 , butwith the buck converter having four instead of three taps;

FIG. 30 shows the power converter of FIG. 26 with a two-phaseswitched-capacitor circuit implementing the switching network and a buckconverter implementing the regulating network;

FIG. 31 shows a power converter with two voltage outputs;

FIG. 32 shows a power converter with three voltage outputs;

FIG. 33 shows a power converter floating above ground;

FIG. 34 is an embodiment in which a 4-switch buck-boost converterimplements the regulating network and a reconfigurable single-phaseasymmetric cascade multiplier implements the switching network;

FIG. 35 is a particular embodiment of the power converter in FIG. 2 inwhich a dual-inductor buck converter implements the regulating networkand a reconfigurable dual-phase asymmetric step-up cascade multiplierimplements the switching network;

FIG. 36 shows an implementation of a power converter in which theswitching network has a separate charge pump embedded within a cascademultiplier;

FIG. 37 shows the embedded charge pump of FIG. 36 ;

FIG. 38 shows the power converter of FIG. 6 in which a buck converterimplements the regulating circuit, a single-phase asymmetric step-upcascade multiplier implements the first switching network, and asingle-phase symmetric step-up cascade multiplier implements the secondswitching network;

FIG. 39 shows the power converter of FIG. 38 in which the buck converteris replaced with a buck-boost converter;

FIG. 40 shows a power converter in which two cascade multipliers withequal numbers of stages implements the switching network;

FIG. 41 shows a power converter in which two cascade multipliers withunequal numbers of stages implements the switching network;

FIG. 42 shows a power converter in which the bulk of the current intothe switching network bypasses the boost converter;

FIG. 43 shows a power converter having floating regulating networks anda grounded switching network; and

FIG. 44 shows an implementation of FIG. 2 in which inductors embedded inthe switching network promote adiabatic charge transfer within theswitching network.

DETAILED DESCRIPTION

FIG. 1 shows a power converter having two stages in series. Depending ondetails of the circuitry within the two stages, and on the operation ofthe controller, the power converter will be a dc-dc converter, an ac-dcconverter, a dc-ac converter, or an ac-ac converter.

Each stage is either a regulating network 16A or a switching network12A. The illustrated power converter separates the function ofvoltage/current transformation from that of regulation. As shown in FIG.1 , it does so by providing a regulating network 16A in series with aswitching network 12A. The two stages can operate at the same frequency,at different frequencies, in phase, and out of phase.

A power source 14 and a load 18A are shown only for clarity. Thesecomponents are not actually part of the power converter. They merelyrepresent the source of the power to be converted, and the ultimateconsumer of that power. Dashed lines between these components and thepower converter indicate that they are optional. Other components shownconnected with dashed lines in this and other figures are likewiseoptional. For example, a dashed wire between the regulating network 16Aand the switching network 12A is also optional.

In FIG. 1 , the power source 14 is a voltage source. However, sincepower is the product of voltage and current, the power source 14 couldjust as easily be a current source. Examples of a suitable power source14 include, but are not limited to, a battery, a solar panel, afuel-cell, and a power supply.

The power source 14 need not deliver a constant stream of power. Infact, if it did, the power converter would not be nearly as necessary.After all, among the tasks of a power converter is to deliver a constantstream of power with specific characteristics to the load 18Anotwithstanding variations in the power stream provided by the powersource 14. The power source 14 is merely a source of power, orequivalently, since power is the time-derivative of energy, a source ofenergy.

The load 18A can be any type of electrical load. What is essential, isthat it be a net energy consumer. Examples of a load 18A include amicroprocessor, LED, RF PA, or a DSP. In fact, the load 18A might evenbe another power converter.

The arrows shown in the figure represent power flow, and not magnitudeof power flow. Hence, each stage can be bidirectional. In such cases, ifthe load 18A supplies power, then the load 18A acts as a power source 14and the power source 14 acts as a load 18A. However, in someembodiments, one or more stages are unidirectional. In addition,embodiments exist in which a stage can be a step-up stage, a step-downstage or a step-up/down stage.

The illustrated regulating network 16A can itself comprise two or moreconstituent regulating circuits operating as a combination in order toregulate some electrical parameter. These regulating circuits can havedifferent voltage ratings and connect to each other in different ways.In some embodiments, the regulating circuits connect in series. Inothers, they connect in parallel, in series-parallel, or inparallel-series.

The regulating circuits that comprise a regulating network 16A can be ofdifferent types. For example, a regulating network 16A may comprise abuck converter in combination with a linear regulator. Examples ofsuitable regulating circuits include a buck converter, a boostconverter, a buck-boost converter, a fly-back converter, a push-pullconverter, a forward converter, a full bridge converter, a half-bridgeconverter, a multi-level converter (buck or boost), a resonantconverter, a Cuk converter, a SEPIC converter, a Zeta converter, and alinear regulator.

Like the regulating network 16A, the switching network 12A can also bemade of a combination of cooperating switching circuits. Theseindividual switching circuits can have different transformation ratios,the same transformation ratios, and different voltage ratings. They canalso be different kinds of switching circuits, such as series parallelor cascade multiplier circuits.

A cascade multipliers includes a switch stack, a phase pump, pumpcapacitors, and, optionally, dc capacitors. The phase pump comprises apair of switches that cooperate to create a pump signal V_(clk). Ingeneral, the two states of the clock are separated by a brief dead-timeto allow transients and the like to decay. In cascade multipliers thatrequire a complement to the clock signal, the phase pump includesanother pair of switches to generate the complement. The switch stack isa series of switches connected between the input and the output of thecascade multiplier.

In those cases in which the switching circuit is a cascade multiplier,it can be asymmetric, symmetric, series-pumped, or parallel-pumped.Additional types of switching circuits include series-parallel switchingcircuits, parallel-series switching circuits, voltage-doubling circuits,and Fibonacci circuits. These constituent switching circuits can connectto each other in series, in parallel, in series-parallel, or inparallel-series. Some configurations of the illustrated power converterpermit adiabatic charge transfer into or out of a capacitor in theswitching network 12A.

Other configurations feature a reconfigurable switching network 12A thattransitions between two or more states in the course of transferringcharge. This charge transfer depends on the voltage across thecapacitor's terminals. Reconfiguring the switching network 12A involvescausing switches in the network to change state to cause this voltage tochange. Reconfiguration can occur, for example, when the voltage orcurrent transformation between the ports of the switching network 12A isto be changed.

FIG. 1 also shows a controller that controls the operation of one ormore stages in the power converter. The controller operates in responseto an I/O signal and a clock signal. In some embodiments, the I/O signalis a digital communication signal. The clock signal can be a signal froma clock or it can be a signal from some analog reference. This might bea signal set by the user. Alternatively, another subsystem sets thissignal and sends it to the power converter.

In some embodiments, the controller receives multiple sensor inputs fromthe power converter and provides control signals along first and secondpaths P1, P2. Examples of sensor signals provided to the sensor inputsare V_(O), V_(X), V_(IN), I_(IN), I_(X) and I₀. Among the foregoingsensor inputs, the negative terminals of V_(IN), V_(X), and V_(O) can beat ground, above ground, or below ground depending upon theircorresponding regulating circuits and switching networks. In fact, sincevoltage reflects a difference in potential energy between two points,there is nothing particularly special about ground.

The controller's function is to control both the regulating network 16Aand the switching network 12A in an effort to control V_(IN), I_(IN),V_(O), and I_(O). In carrying this out, the controller can use eitherfeed-forward control or feedback control. Feed-forward control involveschoosing an output control signal based on an input, whereas feedbackcontrol involves choosing an output control signal based on an output.

Additional control methods that are applicable include voltage-modecontrol, current-mode control, hysteretic control, PFM control,pulse-skipping control, and ripple-based control. In embodiments thatrely upon voltage-mode control, control can be linear or non-linear. Inembodiments that rely upon current-mode control, current can be based onboth an average value of current or a peak value of current.

It is possible to interconnect the regulating network and the switchingnetwork in a variety of ways. FIGS. 1, 2 5, and 6 show four fundamentalbuilding blocks.

In particular, FIG. 1 shows a switching network 12A in series with aregulating network 16A, with the load 18A connected to the switchingnetwork 12A and the power source connected to the regulator network 16A.

FIG. 2 is similar to FIG. 1 but with the power source 14 and load 18Ahaving been swapped. Thus, in FIG. 2 , the power source 14 connects tothe switching network 12A and the load 18A connects to the regulatingnetwork 16A. When power is supplied from the load 18A in the powerconverter shown in FIG. 1 , the result is a power converter that isequivalent to that shown in FIG. 2 .

As mentioned above, it is possible for a switching network 12A tocomprise two or more switching circuits. FIG. shows a particular examplein which a switching network 12A has two switching circuits in series.Each switching circuit is a stage. Assuming the same switched-capacitortopology for each stage, the resulting switching network 12A achieves alarger transformation ratio with the same number of switches andcapacitors. Alternatively, the switching network 12A can achieve thesame transformation ratio, but with fewer switches and capacitors. Onthe other hand, a disadvantage of the power converter shown in FIG. 3 isthat the voltage stresses on the components in at least one of the twostages increases when compared to the single stage case.

FIG. 4 shows an embodiment in which the switching network 12A has firstand second switching circuits connected in series-parallel. The firstswitching circuit achieves a 4:1 transformation ratio and the secondswitching circuit achieves a 3:1 transformation ratio. As a result, anintermediate voltage V_(X) between the regulating network 16A and theswitching network 12A can be any fraction of the output voltage V_(O).In the illustrated power converter, the intermediate voltage V_(X)equals the output voltage V_(O). However, had the first switchingcircuit provided a 4:1 transformation ratio and had the second switchingcircuit provided a 2:1 transformation ratio, then the intermediatevoltage VA would have been larger than the output voltage V_(O).Similarly, had the first switching circuit provided a 4:1 transformationratio and had the second switching circuit provided a 7:2 transformationratio, then the intermediate voltage V_(X) would have been smaller thanthe output voltage V_(O).

FIG. 5 and FIG. 6 show representative three-stage embodiments.

The embodiment of FIG. 5 is similar to that of FIG. 1 , but with theinclusion of an additional regulating network 16B connected to theswitching network 12A. As a result, the switching network 12A is betweentwo regulating networks 16A, 16B.

Conversely, the embodiment of FIG. 6 is similar to that of FIG. 2 , butwith the inclusion of an additional switching network 12B connected tothe regulating network 16A. As a result, the regulating network 16A isbetween two switching networks 12A, 12B.

The four building blocks described above combine in various ways. Forexample, combining the building block in FIG. 5 with that in FIG. 1would result in a power converter in which a first regulating networkconnects to an input of a first switching network, the output of whichconnects to an input of a second regulating network. The output of thissecond regulating network leads to an input of a second switchingnetwork. In the resulting power converter, the power source 14 connectsto an input of the first regulating network and the load 18A connects toan output of the second switching network. FIGS. 7-9 are circuits thatare suitable for modeling the behavior of a switching network 12A. Inparticular, a switching network 12A provides a voltage transformationfrom a first voltage V₁ to a second voltage V₂, with the second voltageV₂ and the first voltage V₁ being related by a ratio of integers.Commonly used ratios are 1:3, 1:2, 1:1, 2:1, and 3:1. The outputresistor R, accounts for voltage drops resulting from finite resistanceof various components. For example, in a switching network 12A that isintended to provide a 1:2 transformation ratio, it would not be uncommonto have an actual transformation ratio of around 1:1.9 instead of theideal 1:2 ratio. The model shown in FIG. 7 differs from that shown inFIGS. 8-9 because the second voltage V₂ can be isolated from the firstvoltage V₁. In addition, it is possible to arbitrarily separate thenegative terminal of the first voltage V₁ and the negative terminal ofthe second voltage V₂. This can be achieved by combining a traditionalswitched-capacitor circuit with a capacitive isolation stage. However,this comes at the price of increased component cost, size, and areduction of efficiency.

The model shown in FIG. 8 differs from that shown in FIG. 7 and FIG. 9because of its common ground. A common ground is desirable in certainapplications. For example, many devices that rely on a battery, such ascell-phones, tablets, and notebooks, use the same ground throughout thedevice.

The model shown in FIG. 9 has a non-common ground. In particular, anoffset voltage V_(off) separates the negative terminal of thetransformer winding on the first voltage V₁ side and the negativeterminal of the transformer winding on the second voltage V₂ side. Theswitched-capacitor topology sets the extent of the offset voltage V₁.This ability to set the offset voltage is advantageous in certainapplications.

FIG. 10 shows a particular implementation of the switching network 12Ain FIG. 1 . In the illustrated embodiment, the switching network 12A isa step-down single-phase asymmetric cascade multiplier having first andsecond sets of one or more switches. These sets of switches will beherein referred to as first and second “switches” 1, 2 for convenience,with the understanding that each can be implemented by multiple switchesoperating in unison. The switching network 12A further includes pumpcapacitors C5-C7, and dc capacitors C1-C4. The pump capacitors C5-C7 arein parallel because their negative terminals all connect to a pumpsignal V_(c1k). The dc capacitors C1-C4 are in parallel because theirnegative terminals connect to ground. In operation, these dc capacitorsC1-C4 store charge from the pump capacitors C5-C7.

During normal operation, the switching network 12A alternates between afirst and second state at a specific frequency and duty cycle, such as50%.

During the first state, the first switch 1 is closed and the secondswitch 2 is open. During the second state, the first switch 1 is openand the second switch 2 is closed. The frequency at which the first andsecond switches 1, 2 both transition between states can be the same asor different from that at which the regulating network 16A switchesbetween states. In cases where these frequencies are the same, they can,but need not be in phase.

The pump capacitors C5-C7 swing up and down as the pump signal V_(clk)alternates between zero volts and the output voltage V_(O). At eachclock cycle, charge moves between a pump capacitor C5-C7 and a dccapacitor C1-C4. In the particular embodiment shown, charge from a dccapacitor C1 makes its way to the last dc capacitor C4 after three clockcycles. Overall, the switching network 12A can be modeled using thecircuit model shown in FIG. 9 with a transformation ratio of 1:1 and anoffset voltage V_(off) of 3V_(O).

In an alternative embodiment, shown in FIG. 11 , the switching network12A is a dual-phase symmetric cascade multiplier having parallel pumpedcapacitors. All switches operate in the same manner already described inconnection with FIG. 10 . Cascade multipliers are described inconnection with the description of FIG. 15 in U.S. Pat. No. 8,860,396,the contents of which are herein incorporated by reference.

A particular feature of the embodiment shown in FIG. 11 is the absenceof dc capacitors inside the cascade multiplier (i.e., capacitors C1, C2are at the input and output terminals). As a result, it is necessary toadd a first switch LA, a second switch 2A, and a first dc capacitor C2.The sole function of the first dc capacitor C2 is to keep the nodevoltage relatively constant. As a result, the capacitance of the firstdc capacitor C2 depends on the amount of charge that is expected to flowin and out of the first dc capacitor C2 as well as the voltage ripplerequirement of the node voltage.

In operation, the first switch LA and the second switch 2A are never ina closed state at the same time. To as great an extent as is possible,the first switch 1A is synchronized with a first switch set 1 so thatwhen the first switch 1A is open, so are all the switches in the firstswitch set 1, and when the first switch 1A is closed, so are all theswitches in the first switch set 1. Similarly, to as great an extent asis possible, the second switch 2A is synchronized with the second switchset 2 so that when the second switch 2A is open, so are all the switchesin a second switch set 2, and when the second switch 2A is closed, soare all the switches in the second switch set 2. As a result, when thepump signal V_(clk) is high, a pump capacitor C8 connects to the firstdc capacitor C2. When the pump signal V_(clk) is low, a pump capacitorC5 is connected to the first dc capacitor C2.

An advantage of the embodiment shown in FIG. 11 is that it becomespossible to create additional dc voltages. For example, it is possibleto create an auxiliary voltage across a second dc capacitor C3. Thisauxiliary voltage thus becomes available to drive other circuits,included such circuits as gate drivers and level shifters, or any othercircuit that would require, during its operation, a voltage that fallsbetween the input voltage V_(IN) and the output voltage V_(O) of thepower converter.

Yet another advantage is the possibility for using charge that is storedin the second dc capacitor C3 to supply a linear regulator, thuscreating a regulated voltage.

In yet another embodiment, shown in FIG. 12 , the switching network 12Afeatures a dual-phase asymmetric cascade multiplier stacked on top ofdual-phase symmetric cascade multiplier with shared phase pump nodes,i.e. the pump signal V_(clk) and its complement. All switches operate inthe same manner already described in connection with FIG. 10 .

Because the nodes are shared in the illustrated embodiment, both cascademultipliers are operated at the same frequency. However, sharing thephase pump is not required.

Additionally, in the particular embodiment shown, the transformationratio of each stage is relatively low. However, there is no specialconstraint on transformation ratio. For example, it would be quitepossible for the dual-phase asymmetric cascade multiplier to have atransformation ratio of 2:1, while the dual-phase symmetric cascademultiplier has a transformation ratio of 10:1.

An advantage of the structure shown in FIG. 12 is that the symmetriccascade multiplier lacks dc capacitors and has fewer switches than acorresponding asymmetric cascade multiplier. Thus, the combination ofthe two as shown in the figure has fewer switches and fewer dccapacitors. On the other hand, the first and second switches S1, S2block twice the voltage as other switches.

FIG. 13A shows a power converter that is a variant of that shown in FIG.10 . In the power converter shown in FIG. 13A, the intermediate voltageV_(X) is applied across the dc capacitors C1, C3 instead of across thedc capacitors C1, C2 as shown in FIG. 10 . The corresponding circuitmodel would be that shown in FIG. 9 , but with a transformation ratio of2:1 and with an offset voltage V_(off) of 2V_(O). In some cases,depending on the nature of the regulating network 16A, the configurationshown in FIG. 13A permits a wider input voltage V_(IN) range, than theconfiguration in FIG. 10 .

FIG. 13B shows a power converter similar to that shown in FIG. 13A, butwith the order of the regulating network 16A and the switching network12A having been reversed. Unlike the configuration shown in FIG. 13A, inwhich the switching network 12A causes the voltage to step down, in theconfiguration shown in FIG. 13B, the switching network 12A causes thevoltage to step up.

FIG. 14A shows a power converter similar to that shown in FIG. 10 withthe exception that the cascade multiplier is a two-phase cascademultiplier instead of a single-phase cascade multiplier. In addition,the pump capacitors C1-C3 are in series instead of in parallel. As aresult, the pump capacitors C1-C3 avoid having an excessively highvoltage across them. On the other hand, the capacitance required ishigher than that required in the switching network 12A shown in FIG. 10. Yet another disadvantage of connecting pump capacitors C1-C3 in seriesis that, when adiabatic charge transfer occurs between such capacitorsduring charge redistribution, the loss associated with suchredistribution is greater than it would be with the switching network12A shown in FIG. 10 .

The cascade multiplier includes a phase pump 32. FIGS. 14B-14C showexamples of suitable phase pumps. Unlike conventional phase pumps, thoseshown in FIGS. 14B-14C include embedded charge pumps. The resultingphase pumps can thus generate pump signals with variable gain. Inparticular, the phase pump 32 shown in FIG. 14B provides voltageattenuation; the phase pump 32 shown in FIG. 14C provides voltage gain.

The phase pump 32 shown in FIG. 14B includes a first pair of switches 1,a second pair of switches 2, a third pair of switches 3, a fourth pairof switches 4, a first pump capacitor C1, and a second pump capacitorC2. In response to receiving an output voltage V_(O), the illustratedphase pump 32 produces a pump signal V_(clk) and its complement.

In operation, the phase pump 32 operates in either a first operatingmode or a second operating mode. In the first operating mode, the pumpsignal V_(clk) alternates between 0 volts and V_(O)/2 volts. In thesecond operating mode, the pump signal V_(clk) alternates between 0volts and V_(O) volts.

Operation in the first mode requires that the phase pump 32 transitionbetween four states according to the following switching pattern inTable 1A:

TABLE 1A State V_(clk) 1 2 3 4 first 0 V OFF ON ON OFF second V_(O)/2OFF ON OFF ON third 0 V OFF ON ON OFF fourth V_(O)/2 ON OFF ON OFF

Operation in the second mode requires that the phase pump 32 transitionbetween two states according to the following switching pattern in Table1B:

TABLE 1B State V_(clk) 1 2 3 4 first 0 V OFF ON ON OFF second V_(O) ONOFF OFF ON

Switching between the first and second mode enables the phase pump 32 tochange the transfer function of the cascade multiplier. For example, ifthe phase pump 32 in FIG. 14A were implemented as shown FIG. 14B, thenin the first mode, the transformation ratio would be 1:2 with an offsetvoltage V_(off) of 2V_(O) volts, but in the second mode thetransformation ratio would be 1:1 with an offset voltage V_(off) of3V_(O) volts.

The phase pump 32 shown in FIG. 14C includes a first set of switches 1,a second set of switches 2, a first pump capacitor C1, and a second pumpcapacitor C2. In response to receiving an output voltage V_(O), theillustrated phase pump 32 produces a pump signal V_(clk) and itscomplement.

In normal operation, the phase pump 32 transitions between a first and asecond state. During the first state, the switches in the first set ofswitches 1 are closed while those in the second set of switches 2 areopened. During the second state, the switches in the first set ofswitches 1 are opened while those in the second set of switches 2 areclosed.

Unlike the phase pump 32 shown in FIG. 14B, which provided analternating signal with a peak below the output voltage V_(O), thisphase pump 32 provides an alternating signal with a peak above theoutput voltage V_(O). Since the peak voltage of the pump signal V_(clk)is twice that of the output voltage V_(O), the phase pump 32 shown inFIG. 14C produces a higher transformation ratio and offset voltageV_(off) than a standard phase pump that uses two pairs of switches.

For example, if the phase pump 32 shown in FIG. 14A were implemented asshown in FIG. 14C, the transformation ratio would be 2:1 with an offsetvoltage V_(off) of 5V_(O) volts compared to 1:1 with an offset voltageV_(off) of 3V_(O) volts for a standard phase pump.

FIG. 15A shows a buck converter that can be used in connection withimplementing a regulating network 16A. The buck converter is similar tothat shown as configuration “A1” in the table in the appendix.

In FIG. 15A, the buck converter has an input terminal, an outputterminal, and a common negative terminal. The common negative terminalis maintained at a common voltage V_(com). The input terminal ismaintained at a first voltage V₁ that differs from the common voltageV_(com) by V₁ volts. The output terminal is maintained at a secondvoltage V₂ that differs from the common voltage by V₂ volts.

The buck converter includes a first switch S1, a second switch S2, aninductor L1, and a driver circuit 20A. The driver circuit 20A receives acontrol signal VR and outputs suitable voltages for controlling thefirst and second switches S1, S2.

It is not actually necessary for the input terminal and the outputterminal to share a common negative terminal. In fact, there are sixpossible configurations of non-isolated regulating circuits that havetwo switches S1, S2, an inductor L1, and four terminals. Two of them arebuck converters. These are shown in FIGS. 15B-15C. Two are buck-boostconverters, which are shown in FIGS. 16A-16B. And the remaining two areboost converters, which are shown in FIGS. 17A-17B. If one is willing tohave four switches S1, S2, S3, S4, additional possibilities emerge, twoof which are shown in FIGS. 18A-18B.

Examples of configurations other than those described above forregulating circuits are shown in the appendix. All of the exemplaryregulating circuits shown in the appendix can be used within theregulating network 16A.

The buck converters shown in FIGS. 15B-15C each have a positive inputterminal and output terminal with their own respective negativeterminal. For ease of discussion, voltages at each terminal of the buckconverters in FIGS. 15B-15C have been assigned names and practicalvalues. Each buck converter has first and second switches S1, S2, aninductor L1, and a drive circuit 20A. The drive circuit 20A receives acontrol signal VR and, in response, outputs voltages that control thefirst and second switches S1, S2. The drive circuit 20A is referenced toa ground potential (i.e., 0 V), which is the negative input terminal inFIG. 15B, and the negative output terminal in FIG. 15C. A level shifteror a boot strap circuit might be required within the driver circuit 20Ato provide appropriate voltage signals to control the first and secondswitches S1, S2.

In operation, the buck converters in FIGS. 15B-15C transition between afirst state and a second state. In the first state, the first switch S1is closed and the second switch S2 is open. In the second state, thefirst switch S1 is open and the second switch S2 is closed. In a powerconverter that uses this regulating network 16A, the voltage +V₂ at thepositive output terminal is lower than the voltage +V₁ at the positiveinput terminal.

In the buck converter shown in FIG. 15B, the negative output terminal,which carries a floating voltage, −V₂, is between two loads 18A, 18B.Meanwhile, the voltage +V₂ is constrained to float between +V₁ and −V₂.In the buck converter shown in FIG. 15C, the negative input terminal,which carries a floating voltage, −V₁, is coupled between two sources14A, 14B. Meanwhile, the voltage +V₂ is constrained to float between +V₁and −V₁.

FIGS. 16A-16B show four-terminal buck-boost converters that are used toimplement the regulating network 16A. In FIG. 16A, the voltage −V₂floats between +V₁ and +V₂. In FIG. 16B, the voltage −V₁ floats between+V₁ and +V₂.

Each buck-boost converter has first and second switches S1, S2, aninductor L1, and a driver circuit 20A that receives a control signal VRand, in response, outputs voltage signals that control the switches S1,S2. The buck-boost converters transition between first and secondstates. In the first state, the first switch S1 is closed and the secondswitch S2 is open. Conversely, in the second state, the first switch S1is open and the second switch S2 is closed. In a power converter thatuses this regulating network 16A, the voltage +V₂ at the positive outputterminal can be higher or lower than the voltage +V₁ at the positiveinput terminal.

FIGS. 17A-17B show boost converters that are used to implement theregulating network 16A. In the boost converter shown in FIG. 17A, thevoltage +V₁ is between +V₂ and −V₂. In the boost converter shown in FIG.17B, the voltage +V₁ is between +V₂ and −V₁.

Each boost converter features first and second switches S1, S2, aninductor L1, and a drive circuit 20A that receives a control signal VRand, in response, outputs voltages suitable for driving the first andsecond switches S1, S2.

In operation, each boost converter transitions between first and secondstates. In the first state, the first switch S1 is closed and the secondswitch S2 is open. In contrast, in the second state, the first switch S1is open and the second switch S2 is closed. In the regulating network16A implemented using the boost converter of FIG. 17A or 17B, thevoltage +V₂ at the positive output terminal is higher than the voltage+V₁ at the positive input terminal.

The regulating network 16A can also be implemented using non-isolatingregulating circuits that have a first switch S1, a second switch S2, athird switch S3, a fourth switch S4, an inductor L1, and four ports. Avariety of configurations are shown in the table in the appendix.

FIGS. 18A-18B show two embodiments of such four-terminal non-invertingbuck-boost converters, each of which can operate in buck mode or boostmode depending on the switch configuration. In buck mode, the regulatorcauses voltage to step down, whereas in boost mode, it causes voltage tostep up.

When operating in buck mode, the first switch S1 is closed and thesecond switch S2 is opened. The remaining switches are then operated totransition between first and second states. In the first state, thethird switch S3 is closed and the fourth switch S4 is open. In thesecond state, the third switch S3 is open and the fourth switch S4 isclosed. In buck mode, +V₂<+V₁ while −V₂<+V₁.

When operating in boost mode, the third switch S3 is closed while thefourth switch S4 is open. The remaining switches are then operated totransition between first and second states. In the first state, thefirst switch S1 is closed and the second switch S2 is opened. In thesecond state, the first switch S1 is open and the second switch S2 isclosed. When operated in boost mode, +V₂>+V₁ while −V₂<+V₁.

A converter along the lines shown in FIGS. 18A-18B is desirable becauseit widens acceptable voltage limits at its terminals. However, it doesso at the cost of increased component cost and size, as well as areduction in efficiency.

A disadvantage of the power converters shown in FIGS. 10-13 is that theoffset voltage V_(off) is fixed to specific fraction of the outputvoltage V_(O). For example, in the power converter shown in FIG. 10 ,the offset voltage V_(off) is 4V_(O). In the power converter shown inFIG. 13 the offset voltage V_(off) is 2V_(O). This is acceptable if theinput voltage V_(IN) or the output voltage V_(O) is constrained over anarrow range. However, if the input voltage V_(IN) or output voltageV_(O) varies over a range that is wider than this narrow range, this canplace the regulating network 16A outside acceptable voltage limitsassociated with its specific implementation. This leads to gaps in theoperating range or over voltage of the switches.

The power converter shown in FIG. 19 overcomes the foregoingdisadvantages. The illustrated power converter is a species of thatshown in FIG. 1 with the switching network 12A implemented using astep-down single-phase asymmetric cascade multiplier having selectionswitches S3-S10 and with the regulating network 16A implemented usingthe regulating circuit shown in FIG. 17A.

The illustrated switching network 12A is an example of a reconfigurableswitched-capacitor network. There are many ways to implement such areconfigurable switched-capacitor network. In fact, in principle, if onecan add any number of switches, there are an infinite number of ways toimplement such a reconfigurable switched-capacitor network. Theregulating network 16A includes first and second active switches 3, 4and an inductor L1. The first and second active switches 3, 4 cyclebetween first and second states at a particular duty cycle andfrequency.

Depending on the upon the required offset voltage V_(off), which is setby input voltage V_(IN) and output voltage V_(O), the offset voltageV_(off) can be set to a fraction of the output voltage V_(O) byselectively enabling and disabling the selection switches S3-S10. Inparticular, the state of each selection switch for the various offsetvoltages V_(off) is shown in Table 2:

TABLE 2 V_(LX) V_(off) S3 S4 S5 S6 S7 S8 S9 S10 V₁/V₂ 3 V₄ ON OFF OFFOFF ON OFF OFF OFF V₂/V₃ 2 V₄ OFF ON OFF OFF OFF ON OFF OFF V₃/V₄ 1 V₄OFF OFF ON OFF OFF OFF ON OFF V₄/gnd 0 V OFF OFF OFF ON OFF OFF OFF ON

Some of the disabled switches (i.e., OFF) will see a higher voltage thanthe active switches in the regulating network 16A.

For example, suppose V₄ equals one volt. Then V₃ equals two volts, V₂equals three volts, and V₄ equals four volts. When operating in the modedescribed on the first line of Table 2, switches S3 and S7 are ON:switch S6 has three volts across it and switch S5 has two volts acrossit, while the active switches 3, 4 only have one volt across them. Ingeneral, the selection switches S3-S10 will either have to have a highervoltage rating than the active switches 3, 4 or they will need to beimplemented as cascaded low-voltage switches.

Another issue with this circuit is that the voltage across the selectionswitches S3-S10 can change polarity. This poses a difficulty because aMOSFET has a parasitic diode in parallel with it, the polarity of whichdepends on where the body contact of the MOSFET is tied. For example, inone embodiment, the first active switch 3 of the regulating network 16Ais a MOSFET having a parasitic diode D1 with its positive terminalconnected to the inductor L1. As a result, the first active switch 3 canonly block a voltage that is higher at the terminal on the output sideof the active switch 3 than the voltage at the inductor side of theactive switch 3. Hence, the selection switches S3-S10 must be able toblock in both directions. One way to do this is to connect two switchesback-to-back with their bodies tied such that the cathodes (negativeterminals) of their corresponding parasitic diodes connect to eachother. Another way to do this is to provide circuitry for changingpolarity of the parasitic diode on the fly, for example by providing abody-snatcher circuit.

The illustrated power converter further includes a disconnect switch S11to protect the low voltage switches in the power converter in the eventof a fault (described in U.S. Pat. No. 8,619,445). The disconnect switchS11 must be a high-voltage switch to achieve this function. Since thisswitch is not routinely operated, it can be made large to reduce itsresistance. However, doing so increases die cost.

In some practices, because a boost converter is practical only when itsduty cycle is between about 5% and 95%, there will be gaps in the spaceof available output voltages.

There are at least two ways to fill these gaps. A first way is to pressthe disconnect switch S11 into service as a linear regulator at theinput of the regulating network 16A. Another way is to place a linearregulator at the output of the regulating network 16A. These both comeat the cost of efficiency. Of the two, placing the linear regulator atthe input is preferable because doing so impairs efficiency less thanplacing the linear regulator at the output.

In principle, it is possible to reconfigure either the regulatingnetwork 16A or the switching network 12A. FIG. 20 shows a circuit inwhich the regulating network 16A is configured to accept or produce,depending upon direction of power transfer, voltages V₁-V₄. Inoperation, the regulating network 16A regulates at least one wire.However, in some embodiments, the regulating network 16A regulates morethan one wire. In those embodiments in which the regulating network 16Aregulates more than one wire, it does so depending upon the value of theinput voltage V_(IN). For example, if the input voltage V_(IN) is low,the regulating network 16A will regulate V₃. If the input voltage V_(IN)is high the regulating network 16A will regulate V₁. This regulatingnetwork 16A reconfigures instead of the switching network 12A.

Instead of using a series of selection switches to present differentvoltages to a regulating network 16A with two output ports, as shownFIG. 19 , it is also quite possible to use a regulating network 16A withmore than two output ports. FIG. 21 shows a power converter having sucha configuration.

The power converter shown in FIG. 21 includes both a reconfigurableswitching network 12A and a reconfigurable switching regulating network16A. Reconfiguration is carried in both cases by reconfigurationswitches. These reconfiguration switches are not involved in actualoperation of the network. Their function is to select different tappoints on the network.

In the power converter shown in FIG. 21 , the reconfigurable switchingnetwork 12A is a single-phase step-down switched-capacitor circuit. Thereconfigurable regulating network 16A is a multi-tap boost converterhaving active switches S1-S5, a cascode switch S6, an inductor L1, and adisconnect switch S11.

During normal operation, only two of the active switches S1-S5 areopening and closing at some specific frequency. The remaining activeswitches are disabled. The boost converter connected to multiple taps onthe switching network 12A. This allows the boost converter to regulatethe voltage difference between two dc capacitors by controlling the timeratio of the enabled active switches. Such control results in aregulated output voltage V_(O). Since the voltage difference between twodc capacitors is the output voltage V_(O), the active switches S1-S5only need to support the output voltage V_(O), which is low voltage. Inthe illustrated examples, each active switch S1-S5 only has to supportone volt. However, the disabled switches would normally have to see ahigher voltage. Selection switches S6-S10 within the switching network12A block this voltage, thereby sparing the disabled active switchesfrom having to endure it.

Table 3 below shows the proper configuration of the switches to achievea particular LX signal V_(LX).

TABLE 3 V_(LX) S1 S2 S3 S4 S5 S6 S7 S8 S9 S10 V₁/V₂ 3 4 OFF OFF OFF OFFON ON OFF OFF V₂/V₃ OFF 3 4 OFF OFF OFF OFF ON ON OFF V₃/V₀ OFF OFF 3 4OFF OFF OFF OFF ON ON V₀/gnd OFF OFF OFF 3 4 ON OFF OFF OFF ON

The disconnect switch S11 is rated to handle the highest voltage. Itsfunction is to disconnect the input from the output. However, it canalso be used during startup, during a power to ground short, and tofunction as a linear regulator in the manner already discussed inconnection with FIG. 19 .

The power converter of FIG. 21 has the same number of switches as thatshown in FIG. 19 . However, it has fewer high-voltage switches. Inparticular, the power converter shown in FIG. 19 has eight high-voltageswitches, whereas the power converter in FIG. 21 has only five. On theother hand, the power converter shown in FIG. 19 has only two activeswitches instead of five for the power converter of FIG. 21 . This meansthat the power converter shown in FIG. 19 requires smaller gate driversand fewer level shifters.

In both cases, selection switches must be able to block in bothdirections. Thus, both the power converter of FIG. 19 and that of FIG.21 include bidirectional switches or body-snatchers.

One disadvantage of the power converter of FIG. 21 is that the activeswitches see both positive and negative polarity. This makesreconfiguring the power converter of FIG. 21 much more difficult andcostly than reconfiguring the power converter shown in FIG. 19 .

FIG. 22 shows a power converter similar to that shown in FIG. 21 , butwith the exception that the switching network 12A is implemented as adual-phase switched-capacitor circuit instead of as a single-phaseswitched-capacitor circuit. In addition, both the pump capacitors C4-C9and dc capacitors C0-C3 are in series. This means that their negativeterminals are not pegged to a common voltage. Despite these differences,the power converters shown in FIGS. 21-22 operate in a similar manner.

The switching network 12A shown in FIG. 22 has more components than thatshown in FIG. 21 . However, each dc capacitor in the dual-phaseswitching network 12A shown in FIG. 22 can have a smaller capacitancethan a corresponding dc capacitor in the single-phase switching network12A shown in FIG. 21 . This is because, in the single-phase switchingnetwork 12A shown in FIG. 21 , the dc capacitors store charge from thepump capacitors. In contrast, in the dual-phase switching network 12A ofFIG. 22 , the dc capacitors only store charge from the boost converterduring the dead-time during which the switching network 12A istransitioning between first and second states.

In the power converters described thus far, adiabatic charging anddischarging of capacitors in the switching network 12A is made possibleby an inductor in the regulating network 16A. However, it is possible toseparate the function of enabling adiabatic charge transfer andregulation by providing a separate magnetic filter. In FIG. 23 , thepower converter of FIG. 1 is modified to include a magnetic filterbetween the load 18A and the switching network 12A. In FIG. 24 , thepower converter of FIG. 1 is modified to include a magnetic filterbetween the power source 14 and the switching network 12A.

It is also possible to incorporate a magnetic filter into the powerconverters shown in FIGS. 5-6 . In fact, as the duty cycle of the secondregulating network 16B shown in FIG. 5 approaches 100%, the secondregulating network 16B effectively becomes a magnetic filter. For aregulating network 12A implemented as shown in FIG. 15C, the secondregulating network 16B can be transformed into a magnetic filter bypermanently closing the first switch S1 and opening the second switchS2.

In some embodiments, the regulating network 16B participates in enablingadiabatic charge transfer even when a magnetic filter is present. Forexample, the magnetic filter may cause a first capacitor to chargeadiabatically while the regulating network 16A causes the same firstcapacitor to discharge adiabatically, or vice versa.

FIG. 25 shows the power converter of FIG. 22 modified to include amagnetic filter between the switching network 12A and the load 18A. Theillustrated magnetic filter comprises an inductor L2 and a capacitor C0.As described in FIGS. 23-24 , the presence of a magnetic filter providesfor adiabatic charging or discharging of capacitors in the switchingnetwork 12A. The magnetic filter constrains the flow of current at theoutput of the switching network 12A thereby reducing the redistributioncurrent among the capacitors in the switching network 12A, and henceloss arising from such redistribution.

The use of a magnetic filter provides another way to span the gap thatarises when, in order to meet a voltage requirement, a regulatingnetwork 16A would have to operate at a duty cycle outside itspermissible range of duty cycles. In embodiments without a magneticfilter, these gaps were filled by using a switch as a linear regulator.However, linear regulators are inefficient.

When a magnetic filter is made available, one can avoid using aninefficient linear regulator to span the gap by chopping the output ofthe switching network 12A and passing it through the magnetic filter toproduce a dc output. In some embodiments, switches in the switchingnetwork 12A carry out the chopping. In other embodiments, an additionalswitch S12 can be added to aid in chopping. Note that elements shownconnected with dotted lines are optional. In those embodiments in whicha buck-boost converter implements the regulating network 16A, neither alinear regulator nor voltage chopping at the switching network 12A isrequired.

Table 4 below summarizes operation of two embodiments, one with and onewithout an additional switch S12. Option 1 of the table shows how theswitches in the switching network 12A transition between first andsecond states to carry out chopping. Option 2 shows how the use of theadditional switch S12 effectively adds a third state between the firstand second states. A benefit of Option 2 is that it avoids having twoseries-connected switches conduct, thus reducing losses. In addition,Option 2 provides for a body diode that can conduct when the switchingnetwork 12A is transiting between states.

TABLE 4 State 3 1A 2A 1B 2B 1 2 S12 OPTION 1 ON ON ON ON OFF OFF —OPTION 2 OFF OFF OFF OFF OFF OFF ON

FIG. 26 shows a power converter that uses a two-phase step-downswitching network 12A connected in series with a step-down regulatingnetwork 16A to limit an output voltage V_(O) to be between two volts andfour volts. The switching network 12A can be of any type. However, agood choice is a cascade multiplier in part because the regulatingnetwork 16A will then permit adiabatic charge transfer betweencapacitors in the switching network 12A. A buck converter having firstand second switches 3, 4, an inductor L1, and a capacitor C1 implementthe regulating network 16A.

During normal operation the regulating network 16A alternates between afirst and second state at a specific frequency and duty cycle, with theduty cycle determining the transformation ratio. During the first state,the first switch 3 closes and the second switch 4 opens. During thefirst switch 3 opens and the second switch 4 closes.

FIG. 27 shows a power converter similar to that shown in FIG. 19 , butwith a wide output voltage V_(O) range instead of a wide input voltageV_(IN) range. Unlike the power converter in FIG. 19 , the powerconverter of FIG. 27 has a regulating network 16A that causes voltage tostep down instead of to step up. In addition, the order of the switchingnetwork 12A and the regulating network 16A are opposite to that shown inFIG. 19 . The resulting configuration enables adiabatic charge transferbetween at least some capacitors in the switching network 12A.

Operation of the power converter in FIG. 27 proceeds along the lines setforth in connection with FIG. 19 . In particular, a controller controlsthe switches according to the timing pattern in Table 2.

Another way to obtain a wide voltage range is to use a step-downswitching network 12A and to implement the regulating network 16A with abuck converter with multiple taps, as shown in FIG. 28 . The switchingnetwork 12A takes an input voltage V_(IN) and provides two voltagerails: a first rail at two volts and a second rail at four volts. Thebuck converter features two modes of operation: a first mode in whichthe LX signal V_(LX) alternates between zero volts and two volts duringa switching cycle, and a second mode in which the LX signal V_(LE)alternates between two volts and four volts in a switching cycle. Thetiming pattern for the switches is set forth in Table 5, with switcheslabeled “ON” being closed during the complete switching cycle and theswitches labeled “OFF” being open during the complete switching cycle.

TABLE 5 V_(LX) 1A 2A 1B 2B 1C 2C 0 V/2 V 3 4 ON OFF ON OFF 2 V/4 V ONOFF 3 4 OFF ON

An advantage of the power converter shown in FIG. 28 is that each switchonly has to support two volts. In addition, the switches are configuredto avoid the need for body-snatcher circuitry. Switches 1B and 2A inparticular have body diodes that point to each other and therefore blockvoltages with any polarity.

To obtain an even wider output range than that given by the powerconverter shown in FIG. 28 , though at the cost of more switches, onecan implement the regulating network 16A with a buck converter havingthree, rather than two, modes of operation. Otherwise, operation issimilar to that described in connection with FIG. 28 .

In FIG. 29 , the switching network 12A maintains three voltage rails attwo volts, four volts, and six volts, respectively. In the firstoperating modes, the LX signal V_(LX) alternates between zero volts andtwo volts during a switching cycle. In the second operating mode, the LXsignal V_(LX) alternates between two volts and four volts during aswitching cycle. In the third operating mode, the LX signal V_(LX)alternates between four volts and six volts during a switching cycle.

The timing diagram for the three operating modes is shown in Table 6:

TABLE 6 V_(LX) 1A 2A 1B 2B 1C 2C 1D 2D 0 V/2 V 3 4 ON OFF OFF OFF ON OFF2 V/4 V ON OFF 3 4 ON OFF OFF OFF 4 V/6 V OFF OFF OFF ON 3 4 OFF ON

Switches labeled “ON” in Table 6 are closed during the completeswitching cycle. Switches labeled “OFF” are open during the completeswitching cycle. As is apparent, each switch only has to support at mosttwo volts. In addition, the switches are properly configured such thatbody snatcher circuits are not required. In particular, switch pair 1B,2A and switch pair 1C, 2B show two body diodes pointing at each other.These switch pairs can therefore block voltages of any polarity.

FIG. 30 shows the power converter of FIG. 26 with a two-phaseswitched-capacitor circuit implementing the switching network 12A and abuck converter implementing the regulating network 16A.

The buck converter, which has two modes of operation, alternates betweenthe switch configurations shown in Table 5. In the first mode, the LXsignal V_(LX) alternates between zero volts and two volts during aswitching cycle. In the second mode, the LX signal V_(LX) alternatesbetween two volts and four volts during a switching cycle. The switcheslabeled “ON” are closed during the complete switching cycle and theswitches labeled “OFF” are open during the complete switching cycle.Each switch only needs to be able support two volts. During normaloperation the switching network 12A alternates between the first andsecond states at a specific frequency and duty cycle. In someembodiments, the duty cycle is 50%.

FIG. 31 shows a power converter with first and second regulators inparallel. This provides first and second outputs. In the particularexample shown, the first output V_(O1) is between four volts and twovolts while the second output V_(O2) is between two volts and zerovolts.

The first regulator includes first and second switches 3, 4 and a firstinductor L1. The second regulator includes third and fourth switches 5,6 and a second inductor L2.

During normal operation the first regulator alternates between a firstand second state at a specific frequency and duty cycle. This duty cycledetermines the transformation ratio. During the first state, the firstswitch 3 is closed and the second switch 4 is opened. During the secondstate, the states are reversed. The second regulator works in the sameway with the third switch 5 replacing the first switch 3 and the fourthswitch 6 replacing the second switch 4. The switching network 12A, thefirst regulator, and the second regulator can operate at the same ordifference frequency and with any phase difference between them.

FIG. 32 shows another embodiment with multiple output voltages. In thiscase, there are three regulating networks 16A, 16B, 16C corresponding tothree output voltages, V_(O1), V_(O2), and V_(O3). Unlike the powerconverter shown in FIG. 31 , these voltages are less limited on thebottom end.

FIG. 33 shows a power converter in which a switching network 12A inseries with a regulating network 16A are floated above ground. Thevoltage upon which the switching network 12A and regulating network 16Afloat comes from a third stage. This third stage can be implementedusing a switch-mode power converter such as a buck converter orswitched-capacitor converter such as a cascade multiplier, or any othertype of power converter. The resulting series connected stages toprovide a higher output voltage than they normally would be able to.Example voltages are provided in the figure. In particular, in FIG. 33 ,the third stage produces a two-volt rail. This effectively boosts theoutput voltage V_(O) by two volts.

An advantage of the configuration shown in FIG. 33 is that only lowvoltages have to be applied across the terminals of the regulatingnetwork 16A. Thus, only low-voltage transistors are needed. These switchvery fast. As a result, one can reduce the size of the inductor.

FIGS. 34-44 show additional embodiments with particular features to bedescribed below. With the exception of a brief dead-time betweentransitions, during which all switches are open to avoid difficultiesthat arise from the finite switching time of a real switch, all switchesin switch set 1 and switch set 2 are always in opposite states, allswitches in switch set 3 and switch set 4 are always in opposite states,and all switches in switch set 5 and switch set 6 are always in oppositestates.

FIG. 34 shows an embodiment in which a 4-switch buck-boost converterimplements the regulating network 16A and a reconfigurable single-phaseasymmetric cascade multiplier implements the switching network 12A. Inthis power converter, the regulating network 16A connects to the middleof the cascade multiplier within the switching network 12A therebyenabling the switching network 12A to cause voltage to either step-up orstep-down.

The regulating network 16A includes an inductor L1, a first switch 3, asecond switch 4, a third switch 5, and a fourth switch 6.

When the regulating network 16A operates in its boost mode, theintermediate voltage V_(X) is higher than the input voltage V_(IN). Inthis mode, the third switch 5 and the fourth switch 6 are active, thefirst switch 3 is closed, and the second switch 4 is open.

Conversely, when the regulating network 16A operates in its buck mode,the intermediate voltage V_(X) is lower than the input voltage V_(IN).In this mode, the first switch 3 and the second switch 4 are active, thethird switch 5 is closed, and the fourth switch 6 is open.

Meanwhile, the switching network 12A includes first and second switchsets 1, 2, four selection switches S1-S4, four dc capacitors C1-C4, andthree pump capacitors C5-C7. The voltages on the four dc capacitorsC1-C4 are 4/2V_(X), 3/2V_(X), 2/2V_(X), and 1/2V_(X), respectively.

In operation, the four selection switches S1-S4 select different dccapacitors C1-C4 within the switching network 12A for presentation tothe load 18A. By properly choreographing the enabling and disabling ofthe selection switches S1-S4 according to a distinct pattern, one canproduce an ac output with a dc offset. This is particularly useful forenvelope tracking when providing power to an RF power amplifier.

FIG. 35 illustrates a variation on the embodiment of FIG. 2 in which adual-inductor buck converter implements the regulating network 16A and areconfigurable dual-phase asymmetric step-up cascade multiplierimplements the switching network 12A.

The regulating network 16A is a dual-inductor buck converter (shown asconfiguration “C1” in the table in the appendix) having a first inductorL1, a second inductor L2, a first capacitor C0, a first switch 3, and asecond switch 4. In some embodiments, the first and second inductors L1,L2 are uncoupled. In others, the first and second inductors L1, L2 arecoupled. These include embodiments that use both positive and negativecoupling.

Unlike in a single-inductor buck converter, the input current into thedual-inductor converter is relatively constant. This results in lowerrms current through the switching network 12A. Both terminals connectedto the switching network 12A draw a relatively constant current. Becauseof this behavior, and because a pump capacitor would always be availableto feed each inductor, the dual-inductor buck converter is best usedwith a full-wave cascade multiplier. It is also possible to use ahalf-wave cascade multiplier. However, in that case, the pump capacitorswould only be feeding the inductors half the time. This requiresproviding high capacitance dc capacitors.

The switching network 12A includes first and second switch sets 1, 2,eight selection switches S1-S8, four dc capacitors C1-C4, and six pumpcapacitors C5-C10.

A third inductor L3 that feeds the switching network 12A promotesadiabatic charge transfer within the switching network 12A. Because itis only filtering a voltage ripple on the capacitors seen at the inputof the switching network 12A, and because it does not have aparticularly large voltage across it, this third inductor L3 has a muchsmaller inductance than those required within the regulating network16A.

Enabling and disabling different selection switches S1-S8 reconfiguresthe switching network 12A, thus enabling one to change the offsetvoltage V_(off) of the switching network 12A. Table 7 shows switchingpatterns used to achieve four different offset voltages.

TABLE 7 V_(off) S1 S2 S3 S4 S5 S6 S7 S8 3 V₄ ON OFF OFF OFF ON OFF OFFOFF 2 V₄ OFF ON OFF OFF OFF ON OFF OFF 1 V₄ OFF OFF ON OFF OFF OFF ONOFF 0 V OFF OFF OFF ON OFF OFF OFF ON

In the particular example shown, the load connected to the output of thepower converter comprises a plurality of light-emitting diodes connectedin series with each other and with the current path through a transistorbiased by a voltage V_(B). This permits control over the LED current,thus enabling the brightness of the LED, which is proportional to theLED current I_(LED), to be controlled. The combination of a powerconverter and a circuit to control the LED current I_(LED), whichamounts to a current sink, is commonly called an LED driver. In mostembodiments, the current sink is somewhat more complicated than a singletransistor. However, the principles illustrated in FIG. 35 areapplicable to such cases as well.

Instead of using a linear regulator to fill the gaps as discussed inreference to FIG. 19 , a current sink along the lines shown in FIG. 35can be used. However, this is also a somewhat inefficient method.

In another embodiment of the power converter of FIG. 1 , which is shownin FIG. 36 , a two-phase boost converter implements the regulatingnetwork 16A while a reconfigurable dual-phase asymmetric step-downcascade multiplier implements the switching network 12A. The particularcascade multiplier shown features an embedded reconfigurable fractionalcharge pump 22.

The two-phase boost converter includes a first inductor L1, a secondinductor L2, a first switch 5, a second switch 6, a third switch 7 and afourth switch 8. A circuit 20A within the regulating network 16Areceives control signals from a controller via a first path P1. Duringnormal operation, the circuit 20A provides first drive signals providedto the first and second switches 5, 6 and second drive signals to thethird and fourth switches 7, 8. The first and second drive signals arein phase quadrature. Controlling the duty cycle of at which the switches4-8 switch regulates the output voltage V_(O).

The switching network 12A includes six selection switches S1-S6, sixpump capacitors C5-C10, four dc capacitors C1-C4, first and secondswitch sets 1, 2, and a circuit 20B that provides drive signals to thefirst and second switch sets 1, 2 and to the six selection switchesS1-S6 based on control signals received along a path P2 from thecontroller.

The reconfigurable fractional charge pump 22 has multiple modes. In theparticular example described herein, the modes are a 1:1 mode and a 3:2mode.

The reconfigurable fractional charge pump 22 enables the parent cascademultiplier in which it is embedded to output half ratios. Table 8 showsthe available transformation ratios (V₄:V_(O)) for the parent cascademultiplier, and both the switch states and the transformation ratio ofthe embedded reconfigurable fractional charge pump 22 that would berequired to achieve those transformation ratios.

TABLE 8 V₄:V₀ S1 S2 S3 S4 S5 S6 22 4.5:1.0 ON OFF OFF ON OFF OFF 3:24.0:1.0 ON OFF OFF ON OFF OFF 1:1 3.5:1.0 OFF ON OFF OFF ON OFF 3:23.0:1.0 OFF ON OFF OFF ON OFF 1:1 2.5:1.0 OFF OFF ON OFF OFF ON 3:22.0:1.0 OFF OFF ON OFF OFF ON 1:1

A particular benefit of providing half ratios is that the resultingpower converter operates without the gaps described in connection withFIG. 36 .

For example, suppose that the output voltage V_(O) is one volt and thatthe input voltage V_(IN) is 3.5 volts. The transformation ratio is 4:1.This requires a duty cycle of 50%, which is well within the permissiblerange for the regulating circuit (i.e., the dual-phase boost converter).

Suppose now that the input voltage V_(IN), drops to 3.05 volts. At thispoint, the required duty cycle at the regulating circuit would drop to5%, which is below the acceptable limit. Ordinarily, this would resultin a gap. But not for the circuit shown in FIG. 36 . This is because oneonly has to switch to a 3.5:1 transformation ratio. This raises therequired duty cycle back up to 65%, which puts it back within thepermissible range of the regulating circuit.

In addition, to eliminating gaps, the ability to provide half ratiosenables the regulating circuit to run with a duty cycle between 25% and75%. This has many benefits, including reducing the rms current and thusboosting the efficiency.

FIG. 37 shows details of the reconfigurable fractional charge pump 22shown in FIG. 36 . The reconfigurable charge pump 22 includes a firstswitch set 3, a second switch set 4, a selection switch S0, a dccapacitor C1, and first and second pump capacitors C2, C3. Thereconfigurable fractional charge pump 22 is connected to have an inputvoltage V₁, an output voltage V₂, and a ground V₃.

The reconfigurable fractional charge pump 22 operates in either a firstmode or a second mode. When operating in the first mode, thereconfigurable fractional charge pump 22 provides a transformation ratioV₁:V₂ of 3:2. When operating in the second mode, the transformationratio is 1:1.

To operate in the first mode, the selection switch S0 opens, and thefirst and second switch sets 3, 4 switch open and close at some specificfrequency. In some embodiments, the switches open and close with a 50%duty cycle. The first switch set 3 and the second switch set 4 arealways in opposite states.

To operate in the second mode, the selection switch S0 closes, and thefirst and second switch sets 3, 4 open.

FIGS. 38-39 show alternative embodiments of the power converter shown inFIG. 6 in which a regulating network 16A is between first and secondswitching networks 12A, 12B. This configuration enables at least oneswitching network 12A, 12B to be adiabatically charged. In addition, theoverall transformation ratio becomes the product of the transformationratios of the first and second switching networks 12A, 12B.

In the embodiment shown in FIG. 38 , a buck converter implements theregulating network 16A, a single-phase asymmetric step-up cascademultiplier implements the first switching network 16A, and asingle-phase symmetric step-up cascade multiplier implements the secondswitching network 16B.

The first switching network 12A includes four dc capacitors C1-C4, threepump capacitors C5-C7, and first and second switches 1, 2. In theconfiguration shown, the first switching network 12A is notadiabatically charged. Hence, it operates with a duty cycle of near 50%.However this is not required because stability is not an issue. Inoperation, the first switching network 12A provides a first voltage V₁that is four times the input voltage V_(IN). However, the intermediatevoltage V_(X) that the regulating network 16A receives from the firstswitching network 12A is twice the input voltage V_(IN).

The regulating network 16A includes a first switch 5, a second switch 6,and an inductor L1. The regulating network 16A controls the duty cycleof the first and second switches 5, 6 to regulate its output voltageV_(O).

The second switching network 12B includes first and second switch sets3, 4, two dc capacitors C11-C12, and three pump capacitors C8-C10.Unlike the first switching network 12A, there is an inductor L1 thatfeeds the second switching network 12B. As a result, the secondswitching network 12B is adiabatically charged.

A dc capacitor C12 connected at the output of the regulating network 16Aimpedes adiabatic operation and is thus optional. This dc capacitor C12is typically added only to maintain stability. As a result, itscapacitance is much smaller than that of other capacitors in thenetwork.

Since the second switching network 12B is adiabatically charged and itstransformation ratio is 1:4, operating it at a duty cycle of 50%promotes stability. The overall output voltage V_(O) of the powerconverter is given by V_(O)=8V_(IN)(D+1), where the duty cycle D isequal to the duty cycle D of the second switch 6 of the regulatingnetwork 16A.

FIG. 39 shows a power converter similar to that shown in FIG. 38 ,except that a buck-boost converter implements the regulating network 16Aand that the first and second switching networks 12A, 12B connect to theregulating network 16A differently. The result is a differentinput-to-output transfer function.

The regulating network 16A includes an inductor L1, a first switch 5,and a second switch 6.

The first and second switching networks 12A, 12B are the same as thoseshown in FIG. 38 . As was the case in FIG. 38 , the second switchingnetwork 12B is adiabatically charged, albeit from a different node. Thesecond switching network 12B includes first and second dc capacitorsC12, C13 that connect to an output of the regulating network 16A. Thesecapacitors impede adiabatic operation, and are thus optional. However,they are often useful for maintaining stability. Preferably, the valuesof their capacitances are much smaller than those used in othercapacitors within the first and second switching networks 12A, 12B.

The differences between the power converter in FIG. 39 and that in FIG.38 enable the power converter shown in FIG. 39 to have an output voltageV_(O) that is either positive or negative. Specifically, for a givenduty cycle D, of the regulating network's second switch 6, the outputvoltage V_(O) is given by where V_(O)=V_(IN)(20D)/(2D−1). Thus, theoutput voltage V_(O) is positive when the duty cycle D exceeds 0.5 andnegative when the duty cycle D falls short of 0.5.

FIGS. 40-43 show embodiments of the power converter shown in FIG. 5 .Each embodiment features a first regulating network 16A and a secondregulating network 16B. In each embodiment, controlling the duty cycle Dof the switches in both the first and second regulating networks 16A,16B permits control over the output voltage V_(O) of the powerconverter. Additionally, in each embodiment, an inductor present in theregulating network 16A permits adiabatic charge transfer betweencapacitors in the switching network 12A. The extent of this adiabaticcharge transfer is one of the differences among the differentconfigurations shown in FIGS. 40-43 .

In the embodiments shown, the first regulating network 16A isimplemented as a step-up converter while the second regulating network16B is implemented as a step-down converter. However, this does not haveto be the case. For example, the order could be reversed, with the firstregulating network 16A causing a step-down in voltage, and the secondregulating network 16B causing a step-up in voltage. Or, both the firstand second regulating networks 16A, 16B could cause a step-up or astep-down in voltage. An advantage of the particular configuration shownin the figures is that if the switching network 12A were areconfigurable switching network, such as that shown in FIG. 19 , theillustrated configuration would permit filling-in the gaps in coverage.

The illustrated embodiments feature a controller to control operation ofswitches in the first and second regulating networks 16A, 16B. Such acontroller can be used to implement a variety of control techniques thatcan be used in connection with controlling the operation of the firstand second regulating networks 16A, 16B. In some embodiments, thecontroller implements feed-forward control over the first regulatingnetwork 16A and feedback control over the second regulating network 16B.In other embodiments, the controller implements feedback over the firstregulating network 16A and feed-forward control over the secondregulating network 16B.

In the embodiment shown in FIG. 40 , the first regulating network 16A isa boost converter and the second regulating network 16B is a buckconverter. The boost converter has a boost-converter inductor L1 andfirst and second boost-converter switches 3, 4. The buck converter has abuck-converter inductor L2 and first and second buck-converter switches5, 6.

The switching network 12A includes first and second switch sets 1, 2,three dc capacitors C1-C3, and six pump capacitors C4-C9 spread acrosstwo symmetric step-up cascade-multipliers. The two cascade multipliersshare a common phase pump and operate 180 degrees out of phase. Theoperation of this switching network 12A is similar to that of adual-phase version, or full-wave, cascade multiplier. The maindistinction arises from separation of the bottom and top of the switchstacks.

Because the stack switches are separate, it is possible to create anintermediate voltage V_(X3) that is a difference between the voltagespresent at the tops of the switch stacks. In the embodiment shown, thevoltage at the top of a first switch stack is 4V_(X), whereas thevoltage at the top of a second switch stack is 3V_(X)+V_(IN). Thus, theintermediate voltage V_(X3) is the difference between these, which isV_(X)−V_(IN).

In the embodiment shown in FIG. 41 , the first regulating network 16A isa three-level boost converter and the second regulating network 16B is abuck converter. The boost converter has a boost-converter inductor L1,four switches 3, 4, 5, 6, and a capacitor C13. The buck converter has abuck-converter inductor L2 and first and second buck-converter switches7, 8.

FIG. 41 shows a power converter similar to that shown in FIG. 40 , butwith the first regulating network 16A being a three-level boostconverter. The three-level boost converter requires less inductance forthe same amount of filtering compared to the boost converter shown inFIG. 40 . In addition, it operates with a reduced voltage stress acrossits switches. However, it also has many more switches and an additionalcapacitor. The second regulating network 16B is again a buck converterlike that shown in FIG. 40 .

The switching network 12A includes first and second switch sets 1, 2,three dc capacitors C1-C3, and nine pump capacitors C4-C12 spread acrosstwo symmetric step-up cascade-multipliers. The two cascade multipliersshare a common phase pump and operate 180 degrees out of phase. Unlikethe switching network 12A shown in FIG. 40 , the two cascade multipliersin FIG. 41 have different numbers of stages. Thus, asymmetry in thenumber of stages results in a third intermediate voltage V_(X3)presented to the second regulating network 16B. In particular, the thirdintermediate voltage, which is 2V_(X), is a difference between a firstvoltage V₁, which is equal to 6V_(X), and a second voltage V₂, which isequal to 4V_(X).

In operation, the three-level boost converter operates in two modes. Ineach mode, the boost converter cycles through first, second, third, andfourth states at a particular frequency. Each state corresponds to aparticular configuration of switches. Table 9A shows the four states inthe first mode, and Table 9B shows the four states in the second mode.

TABLE 9A State V_(LX) 3 4 5 6 first V_(X) ON OFF ON OFF second V_(X)/2OFF ON ON OFF third V_(X) ON OFF ON OFF fourth V_(X)/2 ON OFF OFF ON

TABLE 9B State V_(LX) 3 4 5 6 first 0 V OFF ON OFF ON second V_(X)/2 OFFON ON OFF third 0 V OFF ON OFF ON fourth V_(X)/2 ON OFF OFF ON

Within each mode, the three-level boost converter regulates its outputby controlling its generalized duty cycle. The generalized duty cycle isequal to a first time interval divided by a second time interval. Thefirst time interval is equal to the amount of time the three-level boostconverter spends in either the first state or the third state. Thesecond time interval is the amount of time the three-level boostconverter spends in either the second state or the fourth state.

When the three-level buck converter operates in the first mode, theintermediate voltage V_(X) is greater than two times the input voltageV_(IN). In contrast, when the three-level buck converter operates in thesecond mode, the intermediate voltage V_(X) is less than two times theinput voltage V_(IN).

An advantage over the power converter shown in FIG. 41 over that shownin FIG. 40 is that it is less complex. In addition, losses resultingfrom charge redistribution within the switching network 12A will bereduced because the capacitors within the switching network 12A willenjoy the benefits of adiabatic charge transfer.

The power converter of FIG. 42 combines features of those shown in FIGS.40-41 . As was the case in the power converter shown in FIG. 40 , theswitching network 12A receives both the input voltage V_(IN) and thefirst intermediate voltage V_(X) to produce a third intermediate voltageV_(X3). However, like FIG. 41 , the number of stages is unequal, withthe asymmetry producing the third intermediate voltage V_(X3).

In the switching network 12A shown in FIG. 42 , the first regulatingnetwork 16A no longer drives the phase pump, as was the case in FIGS.40-41 . Thus, the phase pump no longer enjoys the benefit of adiabaticcharge transfer resulting from the intervention of the inductor L1 inthe first regulating network 16A. To make up for this, the switchingnetwork 12A has an additional inductor L3 that promotes adiabatic chargetransfer.

In operation, a first voltage V₁ is equal to V_(X)+5V_(IN), a secondvoltage V₂ is equal to V_(X)+3V_(IN), and a third intermediate voltageV_(X3) is equal to 2V_(IN).

An advantage of the power converter shown in FIG. 42 over that shown inFIG. 41 is that only a fraction of the input current actually passesthrough the three-level boost converter. The bulk of the current insteadbypasses the three-level boost converter and proceeds directly into thephase pump.

Additionally, since the additional inductor L3 only has to promoteadiabatic charge transfer, it can have a smaller inductance that theinductor L1 in the boost converter. This, in turn, reduces resistiveinductor losses.

However, a disadvantage of the power converter shown in FIG. 42 is thatan additional inductor L3 is required. In addition, the cascademultiplier requires more stages to achieve the same voltage gain. Forexample, the first voltage V₁ is equal to 6V_(X) in FIG. 41 while it isequal to V_(X)+5V_(IN) in the circuit shown in FIG. 42 .

FIG. 43 shows a power converter in which a boost converter implementsthe first regulating network 16A, a buck converter implements the secondregulating network 16B, and a dual-phase asymmetric step-up cascademultiple implements the switching network 12A.

The first regulating network 16A includes a first switch 3, a secondswitch 4, and an inductor L1. The second regulating network 16B includesa first switch 5, a second switch 6, and an inductor L2. The switchingnetwork 12A includes a first switch set 1, a second switch set 2, fourdc capacitors C1-C4, and six pump capacitors C5-C10.

In the power converter shown in FIG. 43 , only the switching network 12Ais connected to ground. The first and second regulating networks 16A,16B both float. This reduces the voltage stress across the switches inthe first and second regulating networks 16A, 16B. Unfortunately, thisalso narrows the acceptable input voltage V_(IN) range and outputvoltage V_(O) range. This disadvantage can be overcome by using areconfigurable switching network 12A, but at the added cost of moreswitches.

FIG. 44 shows a particular implementation of the power converter in FIG.2 in which a reconfigurable dual-phase symmetric step-up cascademultiplier implements the switching network 12A and a Zeta converterimplements the regulating network 16A.

The regulating network 16A includes a first inductor L3, a secondinductor L4, a capacitor C10, a first switch 3, and a second switch 4.Depending upon the duty cycle, a Zeta converter can step-up or step-downthe voltage. However, a disadvantage of the Zeta converter is therequirement for more passive components. In addition, a Zeta converteris more difficult to stabilize because of the additional poles and zerointroduced.

The switching network 12A includes first and second switch sets 1, 2;two selection switches S1-S2, three dc capacitors C1-C3, six pumpcapacitors C4-C9, a first inductor L1, and a second inductor L2.

In operation, the switching network 12A transitions between a first modeand a second mode. During the first mode, the first selections switch S1is closed and the second selection switch S2 is open. The intermediatevoltage V_(X) is then V_(IN)/2. During the second mode, the firstselection switch S1 is open and the second selection switch S2 isclosed. In the second mode, the intermediate voltage V_(X) becomes theinput voltage V_(1h).

One way to achieve adiabatic inter-capacitor charge transfer within theswitching network 12A is to place a small inductor in series with thesecond switch S2. However, although this would promote adiabatic chargetransfer during the first mode, it would not do so during the secondmode.

Another way to achieve adiabatic inter-capacitor charge transfer withinthe switching network 12A is to embed the first inductor L1 within thecharge pump and the second inductor L2 in series with the groundterminal of the charge pump.

Preferably the first inductor L1 is embedded at a location that carriesa constant current and that connects to charging and discharging pathsof as many pump capacitors C4-C9 as possible. A suitable location istherefore at the phase pump.

A charge pump typically has two nodes that carry constant current. Asshown in FIG. 44 , the first inductor L1 is at one of these nodes andthe second inductor L2 is at the other. However, only one of theseinductors is actually required to promote adiabatic charge transfer.

Having described the invention, and a preferred embodiment thereof, whatis claimed as new, and secured by Letters Patent is:
 1. An apparatushaving an input port and an output port, the apparatus comprising: acontroller to generate one or more control signals; and a switchingnetwork to include at least a first group of switches and a second groupof switches interconnected with a plurality of capacitors to implementone or more switching patterns via one or more switch configurationsbased, at least in part, on the one or more control signals, the one ormore switching patterns to be implemented to transition the switchingnetwork between at least two states to facilitate an adiabatic chargetransfer within the switching network, wherein the apparatus to includea power path to couple the input port and the output port, wherein theswitching network to be arranged in an electrical configuration with aregulator network to provide power to series-connected LEDs, theelectrical configuration to include a first inductor to be disposed onthe power path to cooperate with the switching network so as to draw arelatively constant current from the switching network, and wherein theadiabatic charge transfer within the switching network to be facilitatedvia a second inductor coupled to the input port.
 2. The apparatus ofclaim 1, wherein the regulating network comprises at least one of thefollowing: a buck converter; a boost converter; a buck-boost converter;a four-terminal non-inverting buck-boost converter; a multi-tap boostconverter; a dual-inductor buck converter; or any combination thereof.3. The apparatus of claim 1, wherein the switching network comprises acascade multiplier.
 4. The apparatus of claim 3, wherein the cascademultiplier comprises an asymmetric step-up full-wave cascade multiplier.5. The apparatus of claim 3, wherein the cascade multiplier comprises areconfigurable dual-phase asymmetric step-up cascade multiplier.
 6. Theapparatus of claim 1, wherein the particular state of the at least twostates to correspond to a particular switch configuration of the one ormore switch configurations.
 7. The apparatus of claim 1, wherein thepower to the series-connected LEDs to be provided via the output port.8. The apparatus of claim 1, the electrical configuration to furtherinclude a third inductor to be disposed on the power path to cooperatewith the switching network so as to draw a relatively constant currentfrom the switching network, wherein the first and the third inductorsare to be positively coupled to each other or to be negatively coupledto each other.
 9. The apparatus of claim 1, wherein the switchingnetwork to include one or more selection switches controllable via theone or more control signals to facilitate outputting a voltage to be afraction of an output voltage.
 10. The apparatus of claim 1, wherein theswitching network to include one or more DC capacitors to store chargefrom the regulating network during a dead time interval to beimplemented via the controller.
 11. The apparatus of claim 1, whereinthe plurality of capacitors to include two or more pump capacitors to bearranged in series.
 12. The apparatus of claim 1, wherein the one ormore control signals to be generated based, at least in part, on avoltage to be provided by the switching network.
 13. A power convertercomprising: a set of capacitors to be interconnected via a plurality ofswitches to form a switched capacitor arrangement coupled to a voltagesource, the plurality of switches to transition between a first and asecond switch configurations, the first and the second switchconfigurations to comprise a full operating cycle of the switchedcapacitor arrangement and to facilitate an adiabatic charge transferwithin the switched capacitor arrangement; a regulating network toreceive a first voltage from the switched capacitor arrangement and togenerate a regulated voltage to power a plurality of series-connectedLEDs; and a current sink electrically coupled in series with theplurality of series-connected LEDs, the current sink to control acurrent flowing through the plurality of series-connected LEDs; whereinthe regulating network comprises coupled inductors that share a magneticcore, wherein the adiabatic charge transfer within the switchedcapacitor arrangement to be facilitated via one or more of the coupledinductors, wherein, during operation of the power converter, the firstswitch configuration to form a first switched capacitor network and thesecond switch configuration to form a second switch capacitor network,and wherein the switched capacitor arrangement to facilitate a voltagetransformation between a first node and a second node of the switchedcapacitor arrangement.
 14. The power converter of claim 13, wherein theswitched capacitor arrangement comprises one or more selection switchesto be controllable to switch between a first and a second configurationso as to generate corresponding first and second offset voltages based,at least in part, on an input voltage from the voltage source.
 15. Thepower converter of claim 13, wherein the regulating network comprises afirst and a second inductors to provide current to the plurality ofseries-connected LEDs.
 16. The power converter of claim 13, wherein theregulating network is to be implemented via a dual-inductor buckconverter.
 17. The power converter of claim 13, wherein the switchedcapacitor arrangement comprises a two-phase step-down switched capacitorarrangement and the regulating network comprises a step-down network.18. The power converter of claim 13, wherein the switched capacitorarrangement comprises a full wave cascade multiplier.
 19. The powerconverter of claim 13, wherein the regulating network is configured tofacilitate the adiabatic charge transfer between two or more capacitorsof the set of capacitors.